Test and measurement instrument including asynchronous time-interleaved digitizer using harmonic mixing

ABSTRACT

A test and measurement instrument including a splitter configured to split an input signal having a particular bandwidth into a plurality of split signals, each split signal including substantially the entire bandwidth of the input signal; a plurality of harmonic mixers, each harmonic mixer configured to mix an associated split signal of the plurality of split signals with an associated harmonic signal to generate an associated mixed signal; and a plurality of digitizers, each digitizer configured to digitize a mixed signal of an associated harmonic mixer of the plurality of harmonic mixers. A first-order harmonic of at least one harmonic signal associated with the harmonic mixers is different from an effective sample rate of at least one of the digitizers.

BACKGROUND

This invention relates to test and measurement instruments and, moreparticularly, to test and measurement instruments including one or moreasynchronous time-interleaved digitizers, which use harmonic mixing forreducing noise.

Useable bandwidths of test and measurement instruments, such as digitaloscilloscopes, can be limited by an analog to digital converter (ADC)used to digitize input signals. The useable bandwidth of an ADC can belimited to the lesser of the analog bandwidth or one half of a maximumsample rate of the ADC. Various techniques have been developed todigitize higher bandwidth signals with existing ADCs.

For example, synchronous time-interleaving can be used to achieve aneffective higher sample rate. Multiple ADCs can sample an input signaloffset in time within a single sample period. The digitized outputs canbe combined together for an effectively multiplied sample rate. However,if the analog bandwidth of the ADCs become the limiting factor, a highbandwidth front end, such as a multi-way interleaved track and holdamplifier is needed to achieve a higher bandwidth.

Conventional track and hold amplifier-based time-interleaved systemscause the track and hold amplifier to be clocked at a sample ratesimilar to or slower than the ADC channel bandwidth so that the ADC willhave sufficient time to settle to the held value. The ADC issynchronously clocked to the track and hold amplifier to digitallycapture each held value. Such a limitation on the track and holdamplifier in turn limits the ADC sample rate. Moreover, to satisfy theNyquist sampling theorem, the ADC sample rate is lowered to less thantwice the bandwidth of the ADC channel. As a result, manytime-interleaved ADC channels are needed to achieve the desiredperformance.

As the number of ADC channels increases, the overall cost and complexityof the system also increases. For instance, the front end chip must nowdrive more ADC channels, including additional ADC circuitry, clockingcircuitry, or the like, to get the overall net sample rate up to asuitable value. The size and complexity of the chip also results inlonger communication paths, and therefore, an increase in parasiticcapacitance, electromagnetic noise, design difficulties, and so forth.

In another technique, sub-bands of an input signal can be downconvertedto a frequency range that can be passed through a lower sample rate ADC.In other words, the wide input bandwidth can be split into multiplelower-bandwidth ADC channels. After digitization, the sub-bands can bedigitally upconverted to the respective original frequency ranges andcombined into a representation of the input signal. One significantdisadvantage of this technique is the inherent noise penalty whendigitizing an arbitrary input signal whose frequency content may berouted to only one ADC channel. The recombined output will containsignal energy from only one ADC, but noise energy from all ADCs, therebydegrading the Signal-to-Noise Ration (SNR).

Accordingly, a need remains for improved devices and methods fordigitizing any frequency input signal by all ADC channels in anasynchronous time-interleaved architecture, thereby avoiding the noisepenalty.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an ADC system for a test and measurementinstrument using harmonic mixing according to an embodiment of theinvention.

FIGS. 2-8 illustrate examples of spectral components of various signalsin the ADC system for the test and measurement instrument of FIG. 1.

FIGS. 9A, 9B and 10-12 are block diagrams of examples of harmonic mixersof FIG. 1.

FIG. 13 is a block diagram of an embodiment of the harmonic mixer ofFIG. 11.

FIG. 14 is a block diagram of an alternative embodiment of the harmonicmixer of FIG. 11.

FIG. 15 is a block diagram of another alternative embodiment of theharmonic mixer of FIG. 11.

FIG. 16 is a block diagram of an alternative harmonic mixer.

DETAILED DESCRIPTION

This disclosure describes embodiments of an ADC system for a test andmeasurement instrument using harmonic mixing.

FIG. 1 is a block diagram of an ADC system for a test and measurementinstrument using harmonic mixing according to an embodiment of theinvention. In this embodiment, the instrument includes a splitter 10configured to split an input signal 12 having a particular frequencyspectrum into multiple split signals 14 and 16, each split signalincluding substantially the entire spectrum of the input signal 12. Asplitter 10 can be any variety of circuitry that can split the inputsignal 12 into multiple signals. For example, the splitter 10 can be aresistive divider. Thus, substantially all frequency components of theinput signal 12 can be present in each split signal 14 and 16. However,depending on the number of paths, harmonic signals used, or the like,the frequency responses for various split signals of a splitter 10 canbe different.

The split signals 14 and 16 are inputs to harmonic mixers 18 and 24,respectively. Harmonic mixer 18 is configured to mix the split signal 14with a harmonic signal 20 to generate a mixed signal 22. Similarly,harmonic mixer 24 is configured to mix the split signal 16 with aharmonic signal 26 to generate a mixed signal 28.

As used herein, a harmonic mixer is a device configured to mix a signalwith multiple harmonics. Although multiplication and/or mixing has beendescribed in connection with harmonic mixing, as will be described infurther detail below, a device that has the effect of multiplying asignal with multiple harmonics can be used as a harmonic mixer.

In some embodiments, the multiple harmonics can include a zero-orderharmonic, or a DC component. For example, in some embodiments, theharmonic signal 20 can be a signal represented by equation (1):

1+2 cos(2πF₁t)   (1)

Here F₁ represents the first-order harmonic and t represents time. Thus,a signal having the form of equation (1) has harmonics at DC and atfrequency F₁.

Harmonic signal 26 can be a signal represented by equation (2)

1−2 cos(2πF₁t)   (2)

Similar to harmonic signal 20, harmonic signal 26 has harmonics at DCand frequency F₁. However, the first-order harmonic at frequency F₁ isout of phase by 180 degrees relative to the similar first-order harmonicin harmonic signal 20.

A digitizer 30 is configured to digitize mixed signal 22. Similarly, adigitizer 32 is configured to digitize mixed signal 28. The digitizers30 and 32 can be any variety of digitizer. Although not illustrated,each digitizer 30 and 32 can have a preamplifier, filter, attenuator,and other analog circuitry as needed. Thus, the mixed signal 22 input tothe digitizer 30, for example, can be amplified, attenuated, orotherwise filtered before digitization.

The digitizers 30 and 32 are configured to operate at an effectivesample rate. In some embodiments, the digitizer 30 can include a singleanalog to digital converter (ADC). However, in other embodiments, thedigitizer 30 can include multiple interleaved ADCs operating at lowersample rates to achieve a higher effective sample rate.

A first-order harmonic of at least one of the harmonic signals 20 and 26is different from an effective sample rate of at least one of thedigitizers 30 and 32. For example, the first-order harmonic F₁ of theharmonic signal 20 could be 34 GHz. A sample rate of the digitizer 30could be 50 GS/s. Thus, the first-order harmonic F₁ is different fromthe effective sample rate.

In some embodiments, the first-order harmonic of a harmonic signal neednot be an integer multiple or sub-multiple of the effective sample rateof the at least one of the digitizers. In other words, in someembodiments, the first-order harmonic of a harmonic signal associatedwith the harmonic mixers is not an integer multiple or sub-multiple ofthe effective sample rate of the at least one of the digitizers.

In some embodiments, the first-order harmonic of a harmonic signal canbe between the effective sample rate of the at least one of thedigitizers and one half of the effective sample rate of the at least oneof the digitizers. In particular, as will be described in further detailbelow, such a frequency allows higher frequency components above and/orbelow the first-order harmonic to be mixed down in frequency to be belowone half of the sample rate of the digitizer 30. Thus, such frequencycomponents can be digitized effectively by the digitizer 30.

It should be understood that all bands of the input signal 12 go throughall paths. In other words, when more than one channel is combined forprocessing a single input signal 12, each channel or path receivessubstantially the entire bandwidth of the input signal 12. As the inputsignal 12 is transmitted through all of the digitizers, the signal tonoise ratio is significantly improved.

A filter 36 can be configured to filter the digitized mixed signal 34from digitizer 30. Similarly, a filter 42 can be configured to filterthe mixed signal 40 from digitizer 32. Harmonic mixers 46 and 52 areconfigured to mix the filtered mixed signals 38 and 44 with harmonicsignals 48 and 54, respectively. In some embodiments, the harmonicsignals 48 and 54 can be substantially similar in frequency and phase tothe corresponding harmonic signals 20 and 26. While the harmonic signals20 and 26 are analog signals, and the harmonic signals 48 and 54 aredigital signals, the scaling factors for these harmonic signals can bethe same or similar to each other. The output signals 50 and 56 arereferred to as remixed signals 50 and 56. A combiner 58 is configured tocombine the remixed signals 50 and 56 into a reconstructed input signal60. In some embodiments, the combiner 58 can implement more than mereaddition of signals. For example, averaging, filtering, scaling, or thelike can be implemented in the combiner 58.

The filters 36 and 42, the harmonic mixers 46 and 52, harmonic signals48 and 54, the combiner 58, and other associated elements can beimplemented digitally. For example, a digital signal processor (DSP),microprocessor, programmable logic device, general purpose processor, orother processing system with appropriate peripheral devices as desiredcan be used to implement the functionality of the processing of thedigitized signals. Any variation between complete integration to fullydiscrete components can be used to implement the functionality.

Some form of synchronization of the harmonic signals 20, 26, 48, and 54is used. For example, the harmonics of the harmonic signals 20 and 26can be locked to a clock related to the digitizers 30 and 32. In anotherexample, the harmonic signal can be digitized. Thus, the first-orderharmonic would be available to synchronize the harmonic signals 48 and54. In another example, out-of-band tones can be added to one or more ofthe mixed signals 22 and 28. Using a first-order harmonic of 34 GHz,19.125 GHz and 21.25 GHz tones, or 9/16 and 10/16 of 34 GHz, can beadded to the mixed signal 22. Since these tones are outside of abandwidth of the filtering eventually established by filter 36, i.e.,approximately 18 GHz depending on the transition band, the tones canhave a substantially negligible effect on the reconstructed signal 60.However, as the tones can be less than a Nyquist frequency, i.e. lessthan 25 GHz for a 50 GS/s sample rate, the tones can be acquired byusing the digitized mixed signal 34 before filtering. Regardless of thetechnique used, a phase and frequency relationship between the harmonicsignals 20 and 26 and the digital harmonic signals 48 and 54 can bemaintained.

FIGS. 2-8 illustrate examples of spectral components of various signalsin the ADC system for the test and measurement instrument of FIG. 1.Referring to FIGS. 1 and 2, spectrum 100 can be a spectrum of the inputsignal 12 and hence, the split signal 14. Using the above example of theharmonic signal defined in equation (1), a DC component of the splitsignal 14 is passed, as represented by spectrum 100. However, thespectrum 100 in the input signal 12 is also mixed with the first-orderharmonic at frequency F₁. The resulting spectrum 102 is the product ofsuch mixing. Thus, the mixed signal 22 includes components of spectrum100 and spectrum 102. Here, and in other figures, the spectralcomponents are illustrated as separate and overlapping however, theactual spectrum would be the combination of the spectra 100 and 102.

Referring to FIGS. 1 and 3, spectrum 110 similarly represents componentsof the mixed signal 28 due to the mixing of input signal 12 with the DCharmonic of the harmonic signal 26. However, in contrast to FIG. 2, thespectrum 112 has a 180 degree phase difference relative to the spectrum102 of FIG. 2. As described above, the first-order harmonic of theharmonic signal 26 is phase shifted by 180 degrees from the first-orderharmonic of the harmonic signal 20. This 180 degree phase shift in theharmonic signal 26 induces a 180 degree phase shift in the spectrum 112.The 180 degree phase difference is illustrated as a dashed line.

FIGS. 4 and 5 represent the spectrums of the filtered mixed signals 38and 44. In some embodiments, the filtering can be a function of inherentfiltering of the corresponding digitizers 30 and 32, the filters 36 and42, or the like. Although filtering is illustrated in FIG. 1 asoccurring after the digitizers 36 and 42, filtering can be performed inother locations. For example, some filtering can occur prior todigitization. The mixed signals 22 and 28 could be filtered with a lowpass filter having a cutoff frequency near one half of the effectivesample rate of the digitizers 30 and 32. The filtering of filters 36 and42 can add to such inherent and/or induced filtering.

In some embodiments, the net filtering of the mixed signals 22 and 28can result in a frequency response that is substantially complementaryabout one half of a frequency of the first-order harmonic of theharmonic signals 20 and 26. That is, the frequency response at a givenoffset higher than frequency F₁/2 and the frequency response at a givenoffset lower than frequency F₁/2 can add to one. Although one has beenused as an example, other values can be used as desired, such as forscaling of signals. Furthermore, the above example is described as anideal case. That is, the implemented filtering can have differentresponse to account for non-ideal components, calibration, or the like.

In a particular example of the frequency response, using the 34 GHz F₁described above, frequency F₁/2 can be 17 GHz. From DC to 16 GHz thefrequency response can be one. From 16 to 18 GHz, the frequency responsecan linearly change from one to zero, passing through ½ at 17 GHz.

The resulting spectral components in FIG. 4, representing the filteredmixed signal 38 include a lower frequency portion of spectrum 100,illustrated by spectrum 120, and a lower frequency portion of spectrum102, illustrated by spectrum 122. Note that due to the mixing, spectrum122 includes frequency components of a higher sub-band of spectrum 100,albeit reversed in frequency. Similarly, the spectral components 130 and132 of FIG. 5 correspond to the lower frequency components of spectra110 and 112 of FIG. 3. The 180 degree phase relationship of spectrum 112is preserved in spectrum 132.

Accordingly, through the harmonic mixing, two sub-bands of an inputsignal 12 have been digitized even though the span of the sub-bandswould have exceeded a Nyquist bandwidth associated with the digitizers30 and 32. In this embodiment, each mixed signal, whether analog,digital, filtered, or the like, includes components of each sub-band ofthe input signal 12. That is, in this example, each signal from themixed signals 22 and 28 to the filtered digitized mixed signal 38 and 44includes both a low frequency sub-band and a high frequency sub-band ofspectrum 100.

In particular, the sub-bands of the input signal 12 have been frequencyshifted to be within the bandwidth of a baseband sub-band. In someembodiments, each sub-band of the input signal 12 can be frequencyshifted to be within the bandwidth of the single sub-band. However,depending on the number of sub-bands, and the harmonic signals, eachsub-band may not be present in each mixed signal.

FIGS. 6 and 7 represent the spectra of the remixed signals 50 and 56.Referring to FIGS. 1 and 6, the spectrum represents the remixed signal50. As described above the filtered digitized mixed signal 38 can bemixed in the harmonic mixer 46 with the harmonic signal 48 that issubstantially similar in frequency and phase to the harmonic signal 20.Accordingly, the spectra of FIG. 4 are mixed with a DC component and afirst-order harmonic.

Spectra 140 and 142 represent the spectra from mixing the spectra 120and 122 of FIG. 4 with the DC component. Spectrum 144 represents theresult of mixing the spectrum 120 with the first-order harmonic. Spectra146 and 148 represent the mixing of spectrum 122 of FIG. 4 with thefirst-order harmonic.

Similarly, FIG. 7 represents the spectra of the remixed signal 56.Spectra 150 and 152 represent the mixing of the DC component with thespectra of FIG. 5. Spectrum 154 represents the mixing of the first-orderharmonic of the harmonic signal 54 with the spectrum 130 of FIG. 5. Inparticular, as the first-order harmonic of harmonic signal 54 has arelative 180 degree phase shift, the resulting spectrum 154 also has a180 degree phase shift, represented by the dashed line.

Spectrum 132 of FIG. 5 is also mixed with the first-order harmonic ofharmonic signal 54; however, the spectrum 132 already had a 180 degreeinduced phase shift. Thus, the additional 180 degree phase shift resultsin an effective 0 degree phase shift, represented by the solid line ofspectra 156 and 158.

FIG. 8 illustrates a spectrum 160 of the reconstructed input signal 60of FIG. 1. Spectra 162 and 164 represent the component sub-bands formingthe spectrum 160. Spectrum 166 represents an additional sideband fromthe mixing described with respect to FIGS. 6 and 7. In this embodiment,spectrum 166 can be filtered out; however, in other embodimentssub-bands can extend beyond the first-order harmonic frequency F₁. Insuch an embodiment, spectrum 166, being generated from a lower frequencysub-band, can be eliminated through destructive combination.

Due to the relative phasing of the components of the remixed signals 50and 56, sub-bands in their original frequency range combineconstructively, while sub-bands outside of their original frequencyrange are phased to combine destructively. Referring to FIGS. 6-8, whencombined, spectra 140 and 150 combine constructively, resulting inspectrum 162. Spectra 142 and 152 combine destructively as the spectraare out of phase by 180 degrees. Thus, of the spectra within thebaseband sub-band, the remaining sub-band is the original sub-band.

Similarly, for the sub-band from approximately F₁/2 to F₁, spectra 146and 156 combine constructively into spectrum 164, while spectra 144 and154 combine destructively. Spectra 148 and 158 combine constructivelyinto spectrum 166; however, spectrum 166 can be filtered out as it isbeyond the expected input frequency range which in this case is aboutless than frequency F₁.

As illustrated by spectra 162 and 164, a transition occurs aroundfrequency F₁/2. This transition is the result of the filtering describedabove in reference to FIGS. 4 and 5. In particular, the slopes ofspectrum 162 and spectrum 164 are complementary. Thus, when thefrequency components of the spectrums 162 and 164 are combined, theresulting portion of the spectrum 160 substantially matches the originalfrequency spectrum.

Accordingly, by mixing the input signal 12 with various harmonicsignals, sub-bands of the input signal 12 can be passed through thelower bandwidth of a digitizer. Although the mixed signals includedoverlapping sub-bands, because of the phasing of the harmonic signals,the sub-bands combine constructively and destructively when combined asdescribed above to create a substantially accurate representation of theinput signal 12.

FIGS. 9-12 are block diagrams of examples of harmonic mixers of FIG. 1.In some embodiments, a mixer can be used to mix the split signals 14 and16 with the respective harmonic signals 20 and 26. A mixer that can passDC and baseband signals on all ports can be used as a harmonic mixer.

FIGS. 9A and 9B illustrate examples of a harmonic mixer, which canrepresent any one or more of the harmonic mixers 18, 24, 46, and/or 52discussed above. FIG. 9A illustrates a 2-way time-interleaving switch.FIG. 9B illustrates an N-way time-interleaving switch.

In these embodiments, switches 180 and/or 181 are configured to receivean input signal 182. When using the 2-way switch 180, the input signal182 is switched to outputs 184 and 186 in response to a control signal188. When using the N-way switch 181, the input signal 182 is switchedto the outputs 184, 186, on through to the Nth output 187, in responseto the control signal 188. For example, the switch 181 can be athree-throw switch, a four-throw switch, etc., up to an N-throw switch,which causes the input signal 182 to spend 1/Nth of its time at eachpoint or output. As further paths and sub-bands are added, the harmonicsof the harmonic signals can be appropriately phased. In someembodiments, the relative phase shifts of the harmonic signals can bespaced in phase by time shifts of one period divided by the number ofsub-bands.

As the pulses get shorter compared to the overall clock cycle, theharmonic content gets richer. For instance, for a two-way or a three-wayswitch, the zero-order harmonic (DC) and the first-order harmonic areused. For a four-way or five-way switch, the zero-order harmonic, thefirst-order harmonic, and a second-order harmonic can be used. For asix-way or seven-way switch, the zero-order harmonic, the first-orderharmonic, a second-order harmonic, and a third-order harmonic can beused. As N increases, the pulses get narrower, thereby generating thericher harmonic content. The control signal 188 can be a signal having afundamental frequency of the first-order harmonic, or other suitableharmonic frequency, described above.

All bands of the input signal 182 go through all paths, i.e., to each ofthe outputs paths (e.g., 184, 186, through the Nth output 187).

For example, referring to switch 180, the control signal 188 can be asquare wave with a fundamental frequency of 34 GHz. As a result of theswitching, output 184 will receive the input signal 182 during onehalf-cycle of the control signal and will be approximately zero duringthe opposite half-cycle. In effect, the output 184 is the input signal182 multiplied by a square wave oscillating between zero and one at 34GHz. Such a square wave can be represented by equation (3).

$\begin{matrix}{0.5 + {\frac{2}{\pi}{\sin \left( {2\pi \; F_{1}t} \right)}} + {\frac{2}{3\pi}{\sin \left( {6\pi \; F_{1}t} \right)}} + \ldots} & (3)\end{matrix}$

Equation (3) is the Taylor series expansion of such a square wave. TheDC and first two harmonics are listed. Here F₁ is 34 GHz. Although themagnitudes of the components are different, equations (1) and (3)include similar harmonics.

Output 186 is similar to output 184; however, the time period over whichthe input signal 182 is routed to the output 186 is inverted relative tooutput 184. The effect is again similar to multiplying the input signal182 with a square wave defined by equation (4).

$\begin{matrix}{0.5 - {\frac{2}{\pi}{\sin \left( {2\pi \; F_{1}t} \right)}} - {\frac{2}{3\pi}{\sin \left( {6\pi \; F_{1}t} \right)}} + \ldots} & (4)\end{matrix}$

Similar to equation (3), equation (4) is similar to the harmonic signaldescribed in equation (2) above. Thus, the multiplication effect of theswitching of the switch 180 is substantially similar to the mixing of asplit signal with the harmonic signal described above. In addition, inthis example, the switch can act as both the splitter 10 and harmonicmixers 18 and 24. However, in other embodiments, the switch 180 could bea single pole single throw switch and act as a single harmonic mixer.

Although the relative magnitudes of the DC component and the first-orderharmonic are different, such imbalance can be corrected through acompensation filter in the appropriate path. For example, the sub-banddescribed above between frequency F₁/2 and frequency F₁ can have adifferent gain applied during recombination in the combiner 58 than abaseband sub-band.

In addition, equations (3) and (4) above also list third-orderharmonics. In some embodiments, the third-order harmonics may bedesired. However, if not, the effect of such harmonics can becompensated with appropriate filtering. For example, the input signal 12can be filtered to remove frequency components above frequency F₁. Thus,such frequency components would not be present to mix with a frequencyat 3*F₁. Moreover, filtering before a digitizer can remove any higherorder frequency components that may otherwise affect the digitizedsignal due to aliasing.

In the event of interleaving errors due to analog mismatch, hardwareadjustments can be made for mixing clock amplitude and phase. Theadjustments can then be calibrated to minimize interleave mismatchspurs. Alternatively, or in addition to the above approach, hardwaremismatches can be characterized, and a linear, time-varying correctionfilter can be used to cancel the interleave spurs. Further, in somecases, the switches might not always operate perfectly. For example, anerrant switch might spend more time in one direction than the other,thereby causing a skewed duty cycle. The digital harmonic mixers 46 and52 can be configured to compensate for phase or amplitude errors thatmay be present in the analog harmonic signals 20 and/or 26 by makingsubtle adjustments to the amplitude or phase of the digital harmonicsignals 48 and/or 54.

FIG. 10 is an example of another harmonic mixer. A switching circuit 200is configured to switch two input signals 202 and 204 alternatively tooutputs 208 and 210 in response to the control signal 206. The controlsignal 206 can again be a square wave or other similar signal to enablethe switches of the switching circuit 200 to switch. During onehalf-cycle of the control signal 206, input signal 202 is switched tooutput 208 while input signal 204 is switched to output 210. During theother half-cycle, the input signal 202 is switched to output 210 whileinput signal 204 is switched to output 208.

In some embodiments, the input signal 204 can be an inverted and scaledversion of the input signal 202. The result of such inputs and theswitching described above is a rebalancing of the DC and other harmonicsfrom the levels described above with respect to the switch 180 of FIG.9A. For example, input signal 204 can be a fractional inverted versionof the inputs signal 202. Instead of switching between 1 and 0 with theswitch 180 of FIG. 9A, the effective output of outputs 208 and 210 canbe switching between 1 and (2−π)/(2+π), for example. Thus, the amplitudeand DC level can be adjusted as desired to create the desired balancebetween the harmonics.

FIG. 11 illustrates an alternative example of a harmonic mixer. Theharmonic mixer 170 includes a splitter 172, a mixer 175, and a combiner177. The splitter 172 is configured to split an input signal 171 intosignals 173 and 174. Signal 174 is input to the combiner 177. As signal174 is not mixed with another signal, signal 174 can act as the DCcomponent of a harmonic mixer described above.

Signal 173 is input to the mixer 175. A signal 176 is mixed with thesignal 173. In some embodiments, signal 176 can be a single harmonic,such as the frequency F₁ described above. If additional harmonics aredesired, additional mixers can be provided and the respective outputscombined in combiner 177.

In another embodiment, the signal 176 can include multiple harmonics. Aslong as the bandwidth of the ports of the mixer 175 can accommodate thedesired frequency ranges, a single mixer 175 can be used. However, sincethe DC component of the harmonic signals described above is passed tothe combiner 177 by a different path, the ports of the mixer receivingsignals 173 and 176 need not operate to DC. Accordingly, a wider varietyof mixers may be used. Once the signals 179 and 174 are combined in thecombiner 177, the output signal 178 can be substantially similar to amixed signal described above.

In some embodiments, the splitter 172 can, but need not split the inputsignal 171 symmetrically. For example, a side of the splitter thatoutputs signal 174 may have a bandwidth that is at or above thefiltering cutoff frequency described above. A side of the splitter 172that outputs signal 173 can have a frequency range centered on aharmonic of the signal 176 and a bandwidth of twice or greater of thefiltering cutoff frequency described above. In other words, thefrequency response of the splitter 172 need not be equal for each pathand can be tailored as desired.

FIG. 12 is another example of a harmonic mixer of the general topologyof FIG. 9A. In this embodiment, a harmonic signal 224 can be input to adiode ring 220 similar to a mixer through transformer 225. The inputsignal 222 can be input to a tap of the transformer 225. Accordingly,depending on the harmonic signal 224, the input signal 222 can beswitched between outputs 226 and 228. For example, the harmonic signal224 causes either the left diodes 227 to turn on when the bottom of thetransformer is positive and the top is negative, or the right diodes 229to turn on when the polarity of the transformer is reversed. In thismanner, the input signal 222 is alternately routed to the output 228 andthe output 226. In some embodiments, an additional diode ring could beused to terminate the outputs and/or inject an inverted portion of asub-band of the input signal 222 to achieve a higher gain, compensatefor imbalanced harmonics, or the like, as in the topology of FIG. 10.

In some embodiments, two paths and two overlapping sub-bands areimplemented. However, as mentioned above, any number of paths andsub-bands can be used. In such embodiments, the number of harmonics usedcan be equal to one plus one half of a number of sub-bands, roundeddown, where DC is included as a zero-order harmonic. For example, forthree sub-bands, only two harmonics can be used. Using the abovefrequency ranges as an example, the first-order harmonic can frequencyshift frequencies higher than frequency F₁ to the baseband sub-band. Thefirst-order harmonics of the harmonic signals can be phased with 120degree relative phase shifts.

Accordingly, when a sub-band is in the proper frequency range duringcombination in the combiner 58, the sub-band spectra will have the samephase shift, such as a 0 degree relative phase shift. In contrast, thethree components of a sub-band in the incorrect frequency range wouldoffset in phase from one another by 120 degrees. The resulting spectrawould destructively combine to eliminate the incorrect sub-band. Asfurther paths and sub-bands are added, the harmonics of the harmonicsignals can be appropriately phased. In some embodiments, the relativephase shifts of the harmonic signals can be spaced in phase by timeshifts of one period divided by the number of sub-bands.

Although embodiments have been described above where digitized signalscan be substantially immediately processed, such processing afterdigitization can be deferred as desired. For example, the digitized datafrom digitizers 30 and 32 can be stored in a memory for subsequentprocessing.

The harmonic mixer 170 of FIG. 11 can be implemented with a hardwareconfiguration that allows for the operation from DC to very widebandwidth using off the shelf components.

FIG. 13 shows one embodiment of the harmonic mixer 170 of FIG. 11 usingoff the shelf components. Harmonic mixer 1300 shown in FIG. 13 is a DCto wide bandwidth harmonic mixer. Harmonic mixer 1300 receives an inputsignal 1302. The input signal 1302 is then split in a power divider, orsplitter, 1304 into a first signal on a first path 1306 and a secondsignal on a second path 1308. Each path 1306 and 1308 includes all ofthe frequencies, including DC, that were present in the input signal1302. As described in further detail below, the power divider 1304 maydivide the input signal 1302 into more than two paths.

The first path 1306, also called a frequency translation path, includesa plurality of off-the-shelf components. For example, as seen in FIG.13, the first signal on the first path 1306 travels through anattenuator 1310, a highpass filter 1312, an amplifier 1314, before beingmixed at a mixer 1316. Attenuator 1310 may be, for example, a −3 dBattenuator. Attenuator 1310 provides input isolation and help withimpedance matching over the low band input to highpass filter 1312.Highpass filter 1312 prevents the high band that is mixed down to thelow band from traveling in a reverse direction in the thru path, andappearing at the input of the mixer 1316. Amplifier 1314 increases theamplitude of the first signal prior to being applied to the mixer 1316.The amplifier 1314 needs to only operate over the rage of ½ the minimumlocal oscillator (LO) frequency, not counting DC, up to the maximumfrequency the design is intended to pass.

Mixer 1316 also receives a harmonic signal 1318 from an LO (not shown).The harmonic signal travels through a bandpass filter 3120 to preventother harmonics, such as from a frequency multiplier circuit, fromentering the mixer 1316. This may be a multiband filter so as to onlypass each of the desired input harmonics and nothing else. The harmonicsignal 1318 also passes through a −3 dB attenuator 1322 to provideisolation and help with impedance matching for LO harmonics applied tothe LO input of the mixer in the first path 1306.

The harmonic signal 1318 from the attenuator 1322 is mixed with thefirst signal on the first path 1306 in the mixer 1316. The mixer 1316outputs a mixed signal 1324 which passes through another −3 dBattenuator 1326 and a lowpass filter 1328 before being input to acombiner 1330. The lowpass filter 1328 has a bandwidth greater than orequal to ½ the lowest LO frequency to be used. Lowpass filter 1328prevents LO harmonics that feed through the mixer 1316 from appear inthe final output 1332 of the overall harmonic mixer 1300.

In the second path 1308, also called the 1.0 thru path, the secondsignal passes through a −6 dB attenuator 1334 and a rigid coax delay1336. The attenuator 1334 helps keep the attenuation consistent betweenthe second path 1308 and the first path 1306. The rigid coax delay 1336passes the second signal on the second path 1308 to the power combiner1330. The rigid coax delay 1336 also provides for a correct delay toallow the second signal on the second path 1308 to arrive at thecombiner at the same time as the first signal on the first path 1306.

Power combiner 1330 in this embodiment is a two-way power combiner. Thepower combiner 1330 combines the output from the rigid coax delay 1336on the second path 1308 with the output from the lowpass filter 1328 onthe first path 1306 and outputs an output signal 1332. The powercombiner 1330 covers a bandwidth from DC to ½ the sample rate of thedigitizers. As discussed in more detail below, the power combiner mayalso be an M-way combiner, where M is the number of paths used withinthe harmonic mixer.

FIG. 14 shows an alternative embodiment for a harmonic mixer. Theembodiment shown in FIG. 14 is similar to that shown in FIG. 13 exceptthe harmonic mixer 1400 now includes two amplifiers 1402 and 1404 tobuffer the first signal on the first path 1306 and the second signal onthe second path 1308. Amplifiers 1402 and 1404 on the first path 1306and the second path 1308, respectively, increases the system gain to bezero, such that an input signal of 0 dBm will result in an output signalthat is close to 0 dBm. The amplifiers 1402 and 1404 also buffer thesignals from passing in the reverse direction thru the power combiner1330.

FIG. 15 shows yet another embodiment of a harmonic mixer. In thisconfiguration, amplifier 1404 has been moved to replace the attenuator1334.

FIG. 16 illustrates an embodiment with a three-way power combiner 1600.In this embodiment, power divider 1602 splits the input signal 1302 intothree signals, the first signal on the first path 1306, the secondsignal on the second path 1308 and a third signal on the third path1604. The first and second paths 1306 and 1308 are identical to thoseshown in FIG. 14. Accordingly, these paths are not further discussedwith respect to FIG. 16.

The third path 1604 is identical to the first path 1306, and is alsocalled a second frequency translation path. However, the harmonic signal1618 is different than the harmonic signal 1318. Additional frequencytranslation paths may be added to the harmonic mixer 1300 as desired.

The third path 1604 includes all of the components of the first path1306. That is, the third path 1604 includes attenuator 1610, highpassfilter 1612, amplifier 1614, mixer 1616, harmonic signal 1618, abandpass filter 1620, another attenutator 1622, a mixed signal 1624, athird attenuator 1626, a lowpass filter 1628 and a second amplifier1606.

The ouput signals in FIGS. 13-16 can be calculated using the followingformulae. The output signal of the mixer equals the input signal timesthe local oscillator as shown in equation (5)

IF=RF·LO, where IF is the output signal, RF is in the input signal.  (5)

Equation (5) can be rewritten with the frequencies of the LO as shown inequation (6):

IF=RF·(1.0+H ₁ +H ₂ + . . . H _(M))   (6)

M in equation 6 is the highest number harmonic needed for a multi-wayinterleave configuration.

H₁, H₂, and H_(M) can be written in terms of the first-order mixingfrequency F₁ as shown in equations (7), (8), and (9):

H ₁=2·cos(2·π·F ₁ ·t)   (7)

H ₂=2·cos(4·π·F ₁ ·t)   (8)

H _(M)=2·cos(2M·π·F ₁ ·t)   (9)

The embodiments of FIGS. 13-16 use a standard triple balanced mixer thatdoes not operate to DC. However, the 1.0 term in the LO input harmonicset when multiplied by the input signal passes the input signal directlythru the mixer without frequency translation. Thus, the 1.0 term isimplemented using the power divider, or splitter, 1302 at the input andthe power combiner 1330 at the output. Therefore, this term is appliedat the input although it is not physically present at the input. Theinput signal passes directly thru the harmonic mixer.

The H₁ and H₂, and higher harmonics, are fed as the harmonic signals1318 and 1618. These terms perform the frequency translation whichaliases multiple bands down to the baseband. Thus, these bands areoverlaid on each other and cover a range of DC up to as much as ½ thesame rate of the digitizers that the harmonic mixer outputs will be feedinto.

While the LO input and input signal to the mixers 1316 and 1616 do notneed to operate to DC, the output of the harmonic mixer does need tooperate to DC for 3-way, 4-way, and higher interleave factors. Theoutput, however, does not need to operate to DC for the 2-way interleavedesigns. The harmonic mixer of FIGS. 13-16 operate from DC to thehighest frequency desired at the input signal. The LO input includingthe implied 1.0 term, must operate over essentially the same range. Theoutput must operate from DC up to as much as ½ of the sample rate of thedigitizer the mixer will be fed to.

Additional frequency translation paths can be added, as shown in FIG. 16discussed above. The addition of the frequency translation paths allowfor wider bandwidth implementation by using multiple mixers. Each mixer,for example, mixers 1316 and 1616 in FIG. 16, cover a different portionof the input signal spectrum. One mixer and amplifier of FIG. 16 needsonly operate over the input spectrum range 25 GHz to 50 GHz. The mixerand amplifier for the other path must operate over the 50 GHz to 75 GHzrange. These mixers can be found in the market place, whereas a mixerthat operates from 25 GHz to 100 GHz cannot be found. A fourth path, orthird translation path, can be added (not shown) to cover the range of75 GHz to 100 GHz. The path would have the same configuration at thefirst path 1305 and the third path 1604 as discussed above.

The harmonic mixers in FIGS. 13-16 can also be used in harmonic timeinterleave (HTI) systems, rather than the asynchronous time-interleavedsystems discussed above. In fact, the harmonic mixers of FIGS. 13-16 canbe used in any system that requires operating from DC to a very highbandwidth.

Although particular values have been discussed with respect to FIGS.13-16, the values are shown as examples. Particular gains and losses canbe adjusted based on available parts, cost trade-offs, etc. Likewise,the bandwidth values can be adapted to meet a market need.

Moreover, although the digital filtering, mixing, and combining havebeen described as discrete operations, such operations can be combined,incorporated into other functions, or the like. In addition, as theabove discussion assumed ideal components, additional compensation, canbe introduced into such processing as appropriate to correct fornon-ideal components. Furthermore, when processing the digitizedsignals, changing frequency ranges, mixing, and the like can result in ahigher sample rate to represent such changes. The digitized signals canbe upsampled, interpolated, or the like as appropriate.

Another embodiment includes computer readable code embodied on acomputer readable medium that when executed, causes the computer toperform any of the above-described operations. As used here, a computeris any device that can execute code. Microprocessors, programmable logicdevices, multiprocessor systems, digital signal processors, personalcomputers, or the like are all examples of such a computer. In someembodiments, the computer readable medium can be a tangible computerreadable medium that is configured to store the computer readable codein a non-transitory manner.

Although particular embodiments have been described, it will beappreciated that the principles of the invention are not limited tothose embodiments. Variations and modifications may be made withoutdeparting from the principles of the invention as set forth in thefollowing claims.

What is claimed is:
 1. A harmonic mixer, comprising: an input signalsplitter configured to split an input signal into a first signal on afirst path and a second signal on a second path; a mixer configured tomix the first signal and a harmonic signal and to generate a mixedsignal on the first path; and a combiner configured to combine thesecond signal from the second path and the mixed signal from the firstpath and to generate an output signal.
 2. The harmonic mixer of claim 1,wherein the mixed signal is a first mixed signal and the harmonic signalis a first harmonic signal, the input signal splitter is furtherconfigured to split the input signal into the first signal, the secondsignal, and a third signal on a third path, the harmonic mixer furthercomprises a second mixer configured to mix the third signal and a secondharmonic signal different from the first harmonic signal and to generatea second mixed signal on the third path, and the combiner is furtherconfigured to combine the second signal, the first mixed signal, and thesecond mixed signal.
 3. The harmonic mixer of claim 1, wherein theharmonic signal is a first harmonic signal, and the mixer is furtherconfigured to mix the first signal, the first harmonic signal, and asecond harmonic signal different than the first harmonic signal.
 4. Theharmonic mixer of claim 1, wherein the input signal is not splitsymmetrically by the input signal splitter.
 5. The harmonic mixer ofclaim 1, wherein the first path includes: a first attenuator to receivethe first signal; a highpass filter to receive the first signal from theattenuator; a first amplifier to receive the signal from the highpassfilter; a second attenuator to receive the mixed signal from the mixer;and a lowpass filter outputs the mixed signal to the combiner.
 6. Theharmonic mixer of claim 5, wherein the second path includes: a thirdattenuator, and a rigid coax delay, wherein the second signal passedthrough the third attenuator and the rigid coax delay before beingcombined in the combiner.
 7. The harmonic mixer of claim 5, wherein thefirst path further includes a second amplifier between the lowpassfilter and the combiner
 8. The harmonic mixer of claim 5, where thesecond path includes: a third amplifier, and a rigid coax delay, whereinthe second signal passed through the third amplifier and the rigid coaxdelay before being combined in the combiner.
 9. The harmonic mixer ofclaim 6, wherein the second path further includes a fourth amplifierbetween the rigid coax delay and the combiner.
 10. The harmonic mixer ofclaim 1, wherein the harmonic signal passes through a bandpass filterand an attenuator before being mixed with the first signal.
 11. Theharmonic mixer of claim 2, wherein the third path includes: a firstattenuator to receive the third signal; a highpass filter to receive thethird signal from the attenuator; a first amplifier to receive thesignal from the highpass filter; a second attenuator to receive themixed signal from the mixer; a lowpass filter; and an amplifier toreceive the second mixed signal from the lowpass filter and output thesecond mixed signal to the combiner.
 12. The harmonic mixer of claim 11,wherein the second harmonic signal passes through a bandpass filter andan attenuator before being mixed with the third signal.
 13. A method,comprising: splitting an input signal into a first signal on a firstpath and a second signal on a second path; mixing the first signal and aharmonic signal and generating a mixed signal on the first path; andcombining the second signal from the second path and the mixed signalfrom the first path and generating an output signal.
 14. The method ofclaim 13, wherein the mixed signal is a first mixed signal and theharmonic signal is a first harmonic signal, the method furtherincluding: splitting the input signal into the first signal, the secondsignal, and a third signal on a third path, mixing the third signal anda second harmonic signal different from the first harmonic signal andgenerating a second mixed signal on the third path, and combining thesecond signal, the first mixed signal, and a second mixed signal. 15.The method of claim 13, wherein the harmonic signal is a first harmonicsignal, and the method further includes mixing on the first path thefirst signal, the first harmonic signal, and a second harmonic signaldifferent than the first harmonic signal.